Method and apparatus for generating code sequence in a communication system

ABSTRACT

A method for transmitting a signal by a transmitting side device, the method including combining a first sequence and a second sequence to generate a third sequence in a frequency domain. The second sequence is obtained by cyclic shifting the first sequence. The method further includes transforming the third sequence into a time domain signal; and transmitting the time domain signal to a receiving side device. The third sequence includes information to identify a cell and is periodically transmitted.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is a Continuation of copending application Ser. No.12/298,488, filed on May 11, 2009, which was filed as the National Phaseof PCT/KR2007/002117 on Apr. 30, 2007, which claims the benefit under 35U.S.C. § 119(a) to Patent Application Nos. 10-2006-0039338 filed in theRepublic of Korea on May 1, 2006, 10-2006-0076146 filed in the Republicof Korea on Aug. 11, 2006 and 10-2006-0076813 filed in the Republic ofKorea on Aug. 14, 2006 , all of which are hereby expressly incorporatedby reference into the present application.

TECHNICAL FIELD

The present invention relates to a mobile communication system, and moreparticularly, to a method for generating a code sequence in a mobilecommunication system and a method and apparatus for processing data fortransmission of the code sequence.

BACKGROUND ART

A pilot signal or preamble, which is used in a mobile communicationsystem, means a reference signal used for acquisition of initialsynchronization, cell detection, channel estimation, etc. An orthogonalor quasi-orthogonal code having good correlation properties may be usedas a code sequence which constitutes the preamble.

For example, in case of PI (Portable Internet, Specifications for 2.3GHz band Portable Internet Service-Physical Layer), 127 sequencesexcluding one of which components are all 1 are inserted in a frequencydomain by using 128×128 Hadamard matrix.

A CAZAC (Constant Amplitude Zero Auto-Correlation) sequence is mainlycharacterized in that its amplitude is uniform and autocorrelation isrepresented by a delta function type. However, cross correlation of theCAZAC sequence, although not zero, has a small value. GCL CAZAC sequenceand zadoff CAZAC sequence, which are mainly used as the CAZAC sequences,are very similar to each other and have different orientations in phase.

First, the GCL CAZAC sequence is given by the following Equations 1 and2.

$\begin{matrix}{{c\left( {{k;N},M} \right)} = {{\exp\left( {- \frac{j\;\pi\;{{Mk}\left( {k + 1} \right)}}{N}} \right)}\mspace{14mu}\left( {{in}\mspace{14mu}{case}\mspace{14mu}{where}\mspace{14mu} N\mspace{14mu}{is}\mspace{14mu}{an}\mspace{14mu}{odd}\mspace{14mu}{number}} \right)}} & \left\lbrack {{Equation}\mspace{14mu} 1} \right\rbrack \\{{c\left( {{k;N},M} \right)} = {{\exp\left( {- \frac{j\;\pi\;{Mk}^{2}}{N}} \right)}\mspace{14mu}\left( {{in}\mspace{14mu}{case}\mspace{14mu}{where}\mspace{14mu} N\mspace{14mu}{is}\mspace{14mu}{an}\mspace{14mu}{even}\mspace{14mu}{number}} \right)}} & \left\lbrack {{Equation}\mspace{14mu} 2} \right\rbrack\end{matrix}$

The Zadoff CAZAC sequence can be expressed by the following Equations 3and 4.

$\begin{matrix}{{c\left( {{k;N},M} \right)} = {{\exp\left( \frac{j\;\pi\;{{Mk}\left( {k + 1} \right)}}{N} \right)}\mspace{14mu}\left( {{in}\mspace{14mu}{case}\mspace{14mu}{where}\mspace{14mu} N\mspace{14mu}{is}\mspace{14mu}{an}\mspace{14mu}{odd}\mspace{14mu}{number}} \right)}} & \left\lbrack {{Equation}\mspace{14mu} 3} \right\rbrack \\{{c\left( {{k;N},M} \right)} = {{\exp\left( \frac{j\;\pi\;{Mk}^{2}}{N} \right)}\mspace{14mu}\left( {{in}\mspace{14mu}{case}\mspace{14mu}{where}\mspace{14mu} N\mspace{14mu}{is}\mspace{14mu}{an}\mspace{14mu}{even}\mspace{14mu}{number}} \right)}} & \left\lbrack {{Equation}\mspace{14mu} 4} \right\rbrack\end{matrix}$

In the Equations 1 to 4 , N is a length of the sequence, M is a naturalnumber which is relatively prime to N among natural numbers less than N,and k represents 0, 1, . . . , N.

Binary Hadamard codes or poly-phase CAZAC codes are orthogonal codes,and the number of the binary Hadamard codes or the poly-phase CAZACcodes, which maintains orthogonality, is limited. The number oforthogonal codes having a length of N, which can be obtained by N×NHadamard matrix, is N, and the number of orthogonal codes having alength of N, which can be obtained by CAZAC code is equivalent to thenumber of natural numbers less than N, wherein the natural numbers arerelatively prime to N. [David C. Chu, “Polyphase Codes with GoodPeriodic Correlation Propertie”, Information Theory IEEE Transaction on,vol. 18, issue 4 , pp. 531-532, July, 1972]

For example, the length of one OFDM (Orthogonal Frequency DivisionMultiplexing) symbol in an OFDM system generally has a length ofexponentiation of 2 to expedite FFT (Fast Fourier Transform) and IFFT(Inverse Fast Fourier Transform). In this case, if the sequence isgenerated by the Hadamard code, sequence types equivalent to the totallength may be generated. If the sequence is generated by the CAZAC code,sequence types equivalent to N/2 may be generated. For this reason, aproblem occurs in that there is limitation in the number of the sequencetypes.

SUMMARY OF THE INVENTION

Accordingly, the present invention is directed to a method and apparatusfor generating a code sequence that can be used for acquisition ofinitial synchronization, cell detection, channel estimation, etc. in acommunication system and a method for processing data for transmissionof the code sequence, which substantially obviate one or more problemsdue to limitations and disadvantages of the related art.

An object of the present invention is to provide a method and apparatusfor extending a range of a code sequence that can be used in acommunication system.

Another object of the present invention is to provide a method andapparatus for processing data to generate a code sequence that canenhance efficiency in acquisition of initial synchronization, celldetection, or channel estimation in a communication system.

For example, if a CAZAC sequence is generated, maximum (N−1) types of acode are available, wherein N is a length of the code. According, toincrease the types of the code, it is preferable to increase the lengthof the code. However, the code is actually determined at a short lengthconsidering bandwidth efficiency of a mobile communication system. Totransmit much information at the same code length, a method forgenerating various kinds of codes with the same length is needed.Particularly, a method which is robust to a multi-path widely occurringin a radio channel and can sufficiently use frequency diversity effectis needed.

One example of the present invention discloses a method for generatingvarious kinds of code sequences by combining at least two different codesequences with each other to generate new code sequences. In otherwords, according to one feature of the present invention, at least twocode sequences are combined with each other to generate one codesequence. The following Equation 5 illustrates one example that at leasttwo code sequences are combined with each other to generate one codesequence.

$\begin{matrix}{{d\left( {{k;\tau_{1}},\tau_{2},\ldots\mspace{14mu},\tau_{L},M_{1},M_{2},\ldots\mspace{14mu},M_{L},N} \right)} = {\sum\limits_{i = 1}^{L}\;{\lambda_{i}{c\left( {{{k - \tau_{i}};M_{i}},N} \right)}}}} & \lbrack{Equation5}\rbrack\end{matrix}$

In the above Equation 5, λ_(i) is a power control constant for thei^(th) code sequence, τ_(i) is a delay component of each code sequence,and M_(i) is a sequence key of the i^(th) code sequence.

The at least two code sequences are combined with each other by addingup of respective corresponding elements of the at least two codesequences, so as to generate one code sequence. Alternatively, the atleast two code sequences may serially be concatenated and combined witheach other to generate one code sequence. Examples of a code to whichtechnical features of the present invention can be applied include, butnot limited to, a binary code, a Hadamard code, and a poly-phase CAZACcode.

In one aspect of the present invention, a method for generating a codesequence in a transmitting side of a mobile communication systemincludes combining at least two code sequences with each other, andconverting a code sequence generated by the combining step into a timedomain signal.

In another aspect of the present invention, a transmitting apparatus forgenerating a code sequence in a mobile communication system includes asequence combiner combining at least two code sequences with each other,and a time domain signal conversion module converting a code sequencegenerated by the combiner into a time domain signal.

In still another aspect of the present invention, a method forprocessing data for transmission of code sequence from a transmittingside of a mobile communication system to its receiving side includesmapping a first code sequence to some subcarriers belonging to a wholeband used in the mobile communication system, mapping a second codesequence to the other subcarriers belonging to the whole band, andconverting the first and second code sequences mapped to the subcarriersinto time domain signals.

In further still another aspect of the present invention, a method forprocessing data for transmission of code sequence from a transmittingside of a mobile communication system to its receiving side includesgenerating a new code sequence by serially concatenating at least twocode sequences, mapping the generated code sequence to some subcarriersbelonging to a whole band used in the mobile communication system, andconverting the code sequence mapped to the subcarriers into a timedomain signal.

In further still another aspect of the present invention, a method forprocessing data for transmission of code sequence from a transmittingside of a mobile communication system to its receiving side includesconverting at least one code sequence into a time domain signal,generating at least two different code sequences by performing circularshift for the at least one code sequence converted into the time domainsignal, and combining the at least two code sequences generated by thecircular shift.

In further still another aspect of the present invention, a transmittingapparatus for generating a code sequence for transmission to a receivingside in a communication system includes an IFFT module converting atleast one code sequence into a time domain signal, a circular shiftmodule generating at least two different code sequences by performingcircular shift for the at least one code sequence converted into thetime domain signal, and a combiner combining the at least two codesequences generated by the circular shift.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a view illustrating the preferred embodiment of the presentinvention;

FIGS. 2A and 2B illustrate examples of hardware configuration for thepresent invention;

FIG. 3 is a view illustrating another preferred embodiment of thepresent invention;

FIG. 4 is a view illustrating another preferred embodiment of thepresent invention;

FIG. 5 is a view illustrating another preferred embodiment of thepresent invention;

FIG. 6 is a view illustrating another preferred embodiment of thepresent invention;

FIG. 7 is a view illustrating another preferred embodiment of thepresent invention;

FIGS. 8A and 8B are views illustrating another preferred embodiment ofthe present invention;

FIG. 9 is a view illustrating an example that values that can be used asdelay components are set in accordance with the preferred embodiment ofthe present invention;

FIG. 10 is a view illustrating steps of the preferred embodiment of thepresent invention; and

FIG. 11 illustrates a performance curve according to the preferredembodiment of the present invention.

DETAILED DESCRIPTION OF THE INVENTION

Hereinafter, structures, operations, and other features of the presentinvention will be understood readily by the preferred embodiments of thepresent invention, examples of which are illustrated in the accompanyingdrawings.

FIG. 1 is a view illustrating the preferred embodiment of the presentinvention. Referring to FIG. 1, an L^(th) CAZAC sequence is obtained insuch a manner that an optional CAZAC sequence, for example, a sequencegenerated by any one of the Equations 1 to 4 is circular shifted byτ_(L). In this case, circular shift means that delay occurs in the CAZACsequence to shift the order of elements. For example, when a codesequence consisting of five elements of A, B, C, D, E is circularshifted by τ_(L)=2 , a code sequence having the order of C, D, E, A, Bis generated. Although the CAZAC sequence generated by circular shifthas orthogonality with the original sequence before circular shift, apeak is generated by cross correlation.

The original CAZAC sequences before circular shift, which correspond tothe first CAZAC sequence to the L^(th) CAZAC sequence, may be the samesequences as one another or different sequences. In other words, afterone CAZAC sequence is circular shifted to generate at least twodifferent CAZAC sequences, the generated CAZAC sequences may be combinedwith each other to generate a new sequence. After each of at least twoCAZAC sequences is circular shifted, the two CAZAC sequences may becombined with each other to generate a new sequence. If at least twodifferent CAZAC sequences are combined with each other, they may notundergo circular shift. Combining at least two different CAZAC sequencesmeans combining corresponding elements of the sequences.

FIG. 2A is a block diagram illustrating a transmitting apparatusaccording to the preferred embodiment of the present invention.Referring to FIG. 2A, at least two different code sequences which havenot undergone circular shift are combined with each other to generate anew code sequence. The transmitting apparatus of FIG. 2A includes acombiner 21 and an IFFT module 22. The combiner 21 receives at least twocode sequences and combines respective elements of the code sequenceswith each other. The IFFT module 22 maps the code sequences generated bythe combiner 21 to subcarriers, and performs IFFT to convert the mappedresult into time domain signals. The combiner 21 may generate one codesequence by adding up corresponding elements of the at least two codesequences. Alternatively, the combiner 21 may generate one code sequenceby serially concatenating the at least code sequences with each other.The at least two code sequences could be binary codes, Hadamard codes,or poly-phase CAZAC codes.

In the case that the at least two code sequences are seriallyconcatenated with each other to generate one code sequence, combiningstep by means of the combiner 21 is omitted, any one of the codesequences is mapped to some subcarriers belonging to the whole band, andthe other code sequence is mapped to the other subcarriers belonging tothe whole band. Afterwards, IFFT is performed for the whole codesequences mapped to the subcarriers, whereby the same result can beobtained.

FIG. 2B is a block diagram illustrating another transmitting apparatusaccording to the preferred embodiment of the present invention.

Referring to FIG. 2B, at least two different CAZAC sequences which haveundergone circular shift are combined with each other to generate a newCAZAC sequence. The transmitting apparatus of FIG. 2B includes an IFFTmodule 23, a circular shift module 24, and an adder 25. The IFFT module23 receives at least one CAZAC sequence and converts the CAZAC sequenceinto a time domain signal. The circular shift module 24 circular shiftsthe at least one CAZAC sequence converted into the time domain signal togenerate at least two different CAZAC sequences. The adder 25 adds theat least two different CAZAC sequences output from the circular shiftmodule 24 to each other so as to generate a new CAZAC sequence.

Although the embodiment of FIG. 2B illustrates the example that theCAZAC sequences are circular shifted in a time domain, the CAZACsequences may be circular shifted before IFFT, i.e., in a frequencydomain to generate a plurality of CAZAC sequences. In this case, atleast one CAZAC sequence is circular shifted before IFFT to generate atleast two CAZAC sequences, and the generated CAZAC sequences arecombined with each other to generate a new CAZAC sequence, whereby thegenerated new CAZAC sequence is converted into a time domain signal byIFFT.

In the Equation 5, λ_(i) is a power control constant for the i^(th)CAZAC sequence, and is to allow a new sequence generated by combinationof at least two CAZAC sequences to perform normalization, therebyallowing each element to obtain amplitude of 1. Two methods fornormalization can be considered. The first method is that each elementof the generated sequences is divided by an absolute value of eachelement. The second method is that the overall power of the generatedsequences is adapted to the number N of the elements.

FIG. 3 is a view illustrating another preferred embodiment of thepresent invention. Referring to FIG. 3, at least two code sequences arecombined with each other to generate one code sequence. If the at leasttwo code sequences combined with each other have different lengths, therespective code sequences are combined with each other in such a mannerthat their lengths are adapted to the that of the longest code sequenceto allow the other code sequences except the longest code sequence tohave repeated patterns. Examples of a method for combining at least twocode sequences with each other include a method for adding upcorresponding elements and a method for serially concatenating codesequences having different patterns. The method for adding upcorresponding elements of at least two code sequences to generate onecode sequence can be expressed by the following Equation 6.

$\begin{matrix}{{d\left( {{k;\tau_{1}},\tau_{2},\ldots\mspace{14mu},\tau_{L},M_{1},M_{2},\ldots\mspace{14mu},M_{L},N_{1},N_{2},\ldots\mspace{14mu},N_{L}} \right)} = {\sum\limits_{i = 1}^{L}\;{\lambda_{i}{c\left( {{{k - \tau_{i}};M_{i}},N_{i}} \right)}}}} & \left\lbrack {{Equation}\mspace{14mu} 6} \right\rbrack\end{matrix}$

According to combination which represents the code generated asexpressed by the Equation 5 or 6 , and (τ₁, τ₂, . . . , τ_(L), M₁, M₂, .. . , M_(L)) and code types having diversity of N^(2L) are generated. Inthis case, although the present invention is intended to sufficientlyincrease codes, if many codes are combined with one another, a receivingside may suffer difficulty in finding each code due to crosscorrelation. Particularly, if delay occurs due to circular shift, amethod for stably estimating such delay is needed. To this end,supposing that τ₁=0 is obtained, relative position for delay of eachsequence can be identified. Accordingly, the method can be used even incase that a start position of the code is unclear. In order thatcombination of keys in delay and code which are detected can be mappedwith a coordinate value of 2L dimension one to one, the relativeposition of delay, i.e., a coordinate position of the code key should beidentified to enable such 1:1 mapping. To this end, a limited conditionis preferably required for delay of each code sequence as expressed bythe following Equation 7.τ_(i), ≤τ_(j), i<j  [Equation 7]

According to the condition of the Equation 7 , although the number ofcodes that can be generated is decreased to reach less than N^(2L), theorder of N^(2L) is maintained.

Accordingly, a coordinate value mapped with the code one to one can beidentified, wherein the code is generated by adding at least two codesequences to each other. From the identified coordinate value, variouskinds of information can be transferred or signal can be identified.

FIG. 4 is a view illustrating another preferred embodiment of thepresent invention. Referring to FIG. 4, one sequence code generated bycombination of at least two CAZAC sequences is applied to asynchronization channel (SCH). Although this one sequence code mayequally be applied a random access channel (RACH), a code expressionvalue may differently be interpreted.

The embodiment of FIG. 4 corresponds to the case of L=2 of the codecombination of FIG. 1. In other words, an expression value of the codeis (τ₁, τ₂, M₁, M₂), wherein τ₁=0 is obtained, and M₁ is set as a keyvalue of a code that will be used in common. In other words, two degreesof freedom are decreased from the above limitation, so that the codeexpression value may be expressed simply by (τ₂, M₂). This means thatthe amount of the codes increases through a delay value of the secondcode and a key value of the code. Supposing that every sample delay canoccur, (N−1)×(N−1) code combinations which are a total number ofavailable cases are generated.

In FIG. 4, it is supposed that four synchronization channels (SCH) existin an interval of 20 ms and the same code is transmitted to eachchannel. The synchronization channels (SCH) are used for acquisition oftime synchronization from a receiving signal of a receiving side,acquisition of overall frame synchronization after time synchronization,and acquisition of cell identifier (cell ID) or cell group ID and othersecondary information. This will now be described briefly.

1) Time synchronization: various methods can be used for timesynchronization. For example, a method for transmitting a code to theSCH may be used. According to this method, a transmitting side transmitsa code symbol to all the subcarriers while a receiving side obtainsdirect cross correlation between transmitting and receiving signals.Alternatively, if the transmitting side does not transmit the codesymbol to all the subcarriers but transmit the code symbol at aninterval of two subcarriers or greater, the signals representperiodicity in a time domain. Accordingly, based on this feature, amethod for processing auto correlation of a receiving signal at areceiving side and searching for a peak may be used.

Of the above two methods, the method for transmitting a symbol to allthe subcarriers has better performance. If the code sequence accordingto the present invention is transmitted, it can be applied to the abovetwo cases. If the cross correlation method is used, a common code M₁ isused to obtain a correlation value. Since M₁ and other sequence added toM₁ are separate codes from each other, they are distinguishable fromeach other. If it is difficult to search for the sequences using oneSCH, the SCHs within different superframes (interval of 20 ms) can beadded to adapt synchronization.

2) Frame synchronization: since it is necessary to identify in whatposition each SCH exists, if SCHs are arranged at constant intervals,they are distinguishable through position information of the sequencesincluded in each SCH or in such a manner that secondary-SCH (S-SCH) isadditionally arranged next to the SCHs. If the code sequence accordingto the present invention is used, the position of each SCH can beidentified by delay. In case of the RACH, other information which anaccess terminal desires can be transmitted along with the RACH.

3) Detection of Cell ID: detection of Cell ID should be obtained fromSCH signals. Since information of the code M₁ has been identified duringtime synchronization, M₁ is removed from the receiving signal, and othersignals can be identified easily by using phase differential incrementaldifference between successive code samples of the CAZAC sequence. Incase of the RACH, various kinds of control information can betransmitted along with the RACH.

FIG. 5 is a view illustrating another preferred embodiment of thepresent invention. The embodiment described with reference to FIG. 5 isapplied to the case where a channel structure is a hierarchicalstructure. Referring to FIG. 5, a subframe of 2 ms hierarchicallyincludes two synchronization channels, i.e., a primary synchronizationchannel (P-SCH) and a secondary synchronization channel (S-SCH).

In FIG. 5, the primary synchronization channel (P-SCH) exists to searchfor time synchronization of a symbol, and the same code sequence can beused in every cell. Although a value greater than 1 can be used as avalue of L, the case L=1 is shown in FIG. 5. A code sequence fortransmission of cell information for cell detection at the receivingside is used for the secondary synchronization channel (S-SCH). At thistime, the code generated in accordance with the preferred embodiment ofthe present invention can be used. Preferably, many kinds of codes canbe used in the secondary synchronization channel (S-SCH) if possible. Inthis case, the value of L may be greater than 1. In case of L=1, onecode key (value of M) is associated with the cell, and much more kindsof information can be transferred depending on the delay value. If thevalue of L is greater than 1, information corresponding to L times ofthe case where the value of L is equal to 1 can be transferred.

FIG. 6 is a view illustrating another preferred embodiment of thepresent invention. The embodiment described with reference to FIG. 6 isapplied to the case where a channel structure is a non-hierarchicalstructure.

The code according to the preferred embodiment of the present inventioncan be applied to each of synchronization symbols, and 1 or a valuegreater than 1 can be applied to the value of L. Furthermore,application of the code sequence generated by the Equation 6 enablesthat the code can be embedded over several synchronization channels,wherein combination of a long code sequence and a short code sequencemay depend on the value of L. Referring to FIG. 6, when synchronizationcodes are inserted into the synchronization channels, foursynchronization symbols may have different delays for the same codesequence or the same delay for different code sequence. Alternatively,synchronization symbols may be grouped (the size of the group is greaterthan 1), so that each group may have the same sequence. Accordingly,information provided by the synchronization channels is represented bycombination of the key value M of the code and the delay value, and ischaracterized in that the information amount becomes L times if Lincreases.

As another preferred embodiment of the present invention, cell detectionsequence employing the present invention will be described. Theembodiment described hereinafter illustrates that technical features ofthe present invention are applied to a hierarchical structure and anon-hierarchical structure, which are discussed as structures of thesynchronization channels in 3GPP LTE (Long Term Evolution). Furthermore,although the embodiment based on the GCL CAZAC code will be describedhereinafter, since the Zadoff-Chu CAZAC sequence is different from theGCL code in symbol component of phase, its application will be the sameas that of the GCL code. Likewise, application of other based CAZACsequence will be the same as that of the GLC code.

Furthermore, the embodiment described hereinafter corresponds to thecase L=1 and will be applied to the OFDM system. Accordingly, theembodiment described hereinafter will be based on circular shift afterthe GCL CAZAC code is generated and then undergoes IFFT. In this regard,the embodiment does not follow the expression method of the aboveequation but employs another notation.

First of all, an example of cell ID detection or cell group ID detectionthrough the S-SCH in the hierarchical structure will be described. TheSCH in the hierarchical structure includes P-SCH and S-SCH. The P-SCH isa channel where all the cells transmit the same signal, through whichinitial synchronization can be acquired. The S-SCH transmits informationof cell-specific ID or cell group ID for each cell and also transmitssecondary information required for acquisition of frame synchronizationand other information. The secondary information required foracquisition of frame synchronization and the other information may betransmitted through a broadcast channel (BCH). Examples of amultiplexing method of the P-SCH and the S-SCH include TDM (TimeDivision Multiplexing), FDM (Frequency Division Multiplexing), and CDM(Code Division Multiplexing) depending on time, frequency, and codedomains. The hierarchical structure shown in FIG. 5 corresponds to thecase where multiplexing of the P-SCH and the S-SCH is TDM. Hereinafter,an example of TDM will be described for convenience.

The P-SCH can be provided by insertion of the GCL CAZAC code or anotheroptional code. In this case, the same code for each cell is used.Accordingly, when the code generated by the preferred embodiment of thepresent invention is applied to the hierarchical structure, it ispreferably used for the S-SCH which transmits cell-specific information.

The code for the S-SCH selects different values of M for each cell andtransmits the selected values. In this case, types of codes that can beidentified for each cell are related to the length N of the code, andthe number of the codes is the number of natural numbers, which arerelatively prime to N among natural numbers less than N. For example, inthe 3GPP LTE system, since the number of subcarriers that can be usedfor synchronization channels of 1.25 MHz is 75 , the number of codesthat can be generated by the Equation 1 is 40(M=1,2,4,7,8,11,13,14,16,17,19,22,23,26,28,29,31,32,34,37,38,41,43,44,46,47,49,52,53,56,58,59,61,62,64,67,68,71,73,74) to generate codes with the length of N=75.Accordingly, the number of a total of cells that can be identified is40. To increase the number of code sequences being generated with thesame length, the following method may be considered. In other words, ifthe CAZAC sequence is generated for the case of N=79, a total of 78 codesequences are generated. At this time, if four elements are truncatedfrom each code sequence to adjust the length of the code sequences toN=75, a total of 78 code sequences of N=75 can be generated.

Secondary information by circular shift sequence according to thepreferred embodiment of the present invention can be used as informationof cell ID classification, information of cell group ID classification,information of bandwidth of cell being transmitted, framesynchronization information, information of the number of transmittingantennas, bandwidth information of another channel such as BCH, andinformation of cyclic prefix (CP) length.

For example, if the CAZAC sequence is generated for the case of N=79 asabove, a total of 78 code sequences are generated. At this time, if eachcode sequence is truncated to adjust the length of the code sequences toN=75 and the number of delay components by means of circular shift foreach code sequence is 8, a total of 624(78×8) circular shift sequencescan be obtained. In this case, supposing that 8 circular shift sequencesare used for the P-SCH in the hierarchical structure, 616 circular shiftsequences can be used for the S-SCH. If the circular shift sequences areused for cell ID classification or cell group ID classification, a totalof 616 cells or cell groups can be identified.

If the code sequences according to the preferred embodiment of thepresent invention are used for cell classification, two cases can beconsidered as follows. In the first case, delay components according tocircular shift are used as cell group ID information, and each codesequence is used as cell ID information. In the second case, each codesequence is used as cell group ID information, and delay componentsaccording to circular shift are used as cell ID information.

In the first case, since 8 delay components according to circular shiftwere supposed, 8 cell groups can be identified. Since the number of thecode sequences is 78, 78 cells for each group can be identified. Thiswill now be described in more detail.

-   -   Cell group ID=0 (delay group 0)    -   Code sequence index: 1, 2, 3, . . . , 78 (a total of 78 types)    -   Cell group ID=1 (delay group 1)    -   Code sequence index: 1, 2, 3, . . . , 78 (a total of 78 types)    -   Cell group ID=2 (delay group 2)    -   Code sequence index: 1, 2, 3, . . . , 78 (a total of 78 types)    -   Cell group ID=3 (delay group 3)    -   Code sequence index: 1, 2, 3, . . . , 78 (a total of 78 types)    -   Cell group ID=4 (delay group 4)    -   Code sequence index: 1, 2, 3, . . . , 78 (a total of 78 types)    -   Cell group ID=5 (delay group 5)    -   Code sequence index: 1, 2, 3, . . . , 78 (a total of 78 types)    -   Cell group ID=6 (delay group 6)    -   Code sequence index: 1, 2, 3, . . . , 78 (a total of 78 types)    -   Cell group ID=7 (delay group 7)    -   Code sequence index: 1, 2, 3, . . . , 78 (a total of 78 types)    -   {the number of a total of cell IDs: 624 types}

In the second case, 78 cell groups corresponding to the number of thecode sequences can be identified, and 8 cells for each group can beidentified. In both the first and second cases, a total of 624 (78×8)cells can be identified. Since one code sequence should be used for theP-SCH in the hierarchical structure, a total of 616 (77×8) cells can beidentified.

To facilitate cell planning during network design, a large number ofcell IDs in a physical level are preferably required if possible. Also,although all cell IDs prescribed on the standard during network designmay be used, some of them may be used. Accordingly, a network designercan use the required number of cell IDs for network design throughgrouping. For example, supposing that there are provided a total of 624available cell IDs and that 234 cell IDs are only required during celldesign, three cell group IDs having 78 cell IDs can be used.

As another example, supposing that the number of required cell IDs is512, 64 code sequences are respectively used for 8 cell groups. Thiswill now be described in more detail.

-   -   Cell group ID=0 (delay group 0)    -   Code sequence index: 1, 2, 3, . . . , 64 (a total of 64 types)    -   Cell group ID=1 (delay group 1)    -   Code sequence index: 1, 2, 3, . . . , 64 (a total of 64 types)    -   Cell group ID=2 (delay group 2)    -   Code sequence index: 1, 2, 3, . . . , 64 (a total of 64 types)    -   Cell group ID=3 (delay group 3)    -   Code sequence index: 1, 2, 3, . . . , 64 (a total of 64 types)    -   Cell group ID=4 (delay group 4)    -   Code sequence index: 1, 2, 3, . . . , 64 (a total of 64 types)    -   Cell group ID=5 (delay group 5)    -   Code sequence index: 1, 2, 3, . . . , 64 (a total of 64 types)    -   Cell group ID=6 (delay group 6)    -   Code sequence index: 1, 2, 3, . . . , 64 (a total of 64 types)    -   Cell group ID=7 (delay group 7)    -   Code sequence index: 1, 2, 3, . . . , 64 (a total of 64 types)    -   {the number of a total of cell IDs: 512 types}

FIG. 7 is a view illustrating another preferred embodiment of thepresent invention. In the embodiment of FIG. 7, when a total of 427 cellIDs are required during network design, the number of cell group IDs is7 and the number of code sequences for each group is 61. In FIG. 7,numbers within cells represent examples of cell group IDs within eachcell group. In FIG. 7, although one cell is comprised of one sector, itis apparent that one cell may be applied to other sector structure suchas three sectors and six sectors.

As another example of delay components by means of circular shiftaccording to the preferred embodiment of the present invention, thedelay components by means of circular shift can be used to indicateinformation of the number of transmitting antennas or the length of CP.For example, the number (78) of the code sequences available by theexisting sequences can be used for cell classification or cell group IDclassification, and the number (8) of the delay components by means ofcircular shift can be used to indicate information of the case where thenumber of transmitting antennas is 4 and types of the CP length are 2.

FIGS. 8A and 8B are views illustrating another preferred embodiment ofthe present invention. FIG. 8A is a flow chart illustrating theprocedure of the transmitting side, and FIG. 8B is a flow chartillustrating the procedure of the receiving side.

Referring to FIG. 8A, the transmitting side generates the codesequences, i.e., GCL CAZAC codes (71). The transmitting side may notgenerate the code sequences directly but store previously generated codesequences so as to output them if necessary. The GCL CAZAC codes can begenerated by the following Equation 8.C ^(M)(k)=exp(−jπk(k+1)/N)  [Equation 8]

-   -   k=0,1, . . . , N−1

In the Equation 8, C^(M)(k) is a GCL code of the kth element having aseed value of M. Since N which is a code length corresponds to N=75 ,the equation where N is an odd number has been used.

The transmitting side maps each subcarrier to the generated codesequences (72). In this step, the transmitting side can insert guardsubcarrier and DC subcarrier in accordance with the following Equation9.C _(guard) ^(M) =f(C ^(M))  [Equation 9]

In this Equation 9 , f( ) is a function which maps a complex signalequivalent to used subcarrier to the guard subcarrier and the DCsubcarrier for IFFT modulation.

The transmitting side performs IFFT for the code sequences mapped to thesubcarriers in accordance with the following Equation 10 to generateOFDM symbols of the time domain.

$\begin{matrix}{{{{c^{M}(n)} = {\frac{1}{N_{fft}}{\sum\limits_{k = 0}^{N_{fft} - 1}\;\left( {{C_{guard}^{M}(k)}{\exp\left( {j\frac{2\pi\;{kn}}{N_{fft}}} \right)}} \right)}}}{k = 0},1,2,\ldots\mspace{14mu},{N_{fft} - 1}}{{n = 0},1,2,\ldots\mspace{14mu},{N_{fft} - 1}}} & \left\lbrack {{Equation}\mspace{14mu} 10} \right\rbrack\end{matrix}$

The transmitting side performs circular shift for the IFFT performedsignal in accordance with the following Equation 11.c _(τ) ^(M)(n)=c ^(M)(mod(N _(fft) +n−τ, N _(fft)))  [Equation 11]

In the Equation 11 , mod(a,b) means the remainder obtained by dividing aby b. The Equation 11 means circular shift to the right. Circular shiftto the left can be obtained by the following Equation 12.c _(τ) ^(M)(n)=c ^(M)(mod(N _(fft) +n+τ, N _(fft)))  [Equation 12]

In view of the frequency domain, the signal generated by the Equation 11is converted into a form of a phase value of each subcarrier, which islinearly increased, as expressed by the following Equation 13.

$\begin{matrix}{{C_{r}^{M}(k)} = {{C_{guard}^{M}(k)} \cdot {\exp\left( {{- j}\frac{2\pi\; k\;\tau}{N_{fft}}} \right)}}} & \left\lbrack {{Equation}\mspace{14mu} 13} \right\rbrack\end{matrix}$

Although it has been described above that circular shift is performed inview of digital aspect, the signal after digital to analog (D/A)conversion may undergo circular delay. At this time, the number ofsamples which undergo circular shift is selected considering delayspread of a channel considered in a system which is used.Conventionally, since the CP length is designed considering such delayspread in an OFDM system, a CP interval range (exactness is notnecessarily required) is considered when this step is performed.

FIG. 9 is a view illustrating an example that values that can be used asdelay components are set in accordance with the preferred embodiment ofthe present invention. In FIG. 9, 128-point IFFT is performed so that 8groups for each of 16 samples are grouped if the number of samples ofOFDM symbols except CP is 128. The reason why grouping is performed fordelay is to allow the receiving side to be robust to influence of achannel and exactly receive a delay value when the receiving sidedetects the delay value. Currently, since the maximum delay spreadlength of a typical urban (TU) model which is a channel model consideredby the LTE is 5 us (10 samples), 16 samples which correspond to 8.33 uscover the maximum delay spread length. Also, since the length of a shortCP is 4.69 us, 5.21 us, defining 8.33 us as one group is adequate evenin case that design is performed based on the length of the short CP.The receiving side can robustly estimate a delay value through a marginin spite of interference such as remaining symbol timing offset. Thegrouping method for grouping 8 groups for each of 16 samples withrespect to the 128 symbols is only exemplary, and various numbers ofgroups and various numbers of samples for each group, such as 16 groupsfor each of 8 samples, may be defined.

Delay combination that can be obtained in this embodiment includes 8types such as 0 us (0 sample), 8.330 us (16 samples), 16.66 us (32samples), . . . , 58.3 lus (112 samples). Although it has been describedthat 128-point IFFT is performed, other-point (256, 512, 1024, 1536,2048) IFFT may be performed if 1.25 MHz (75 subcarriers) is used for theSCH and the other band is used for other purpose in the same manner asthe LTE. In either case, the time corresponding to one OFDM symbollength is uniformly maintained at 66.7 us even though the number of thesamples depends on each case.

Hereinafter, a method for detecting a code at a receiving side inaccordance with the preferred embodiment of the present invention willbe described with reference to FIG. 8B.

Referring to FIG. 8B, the receiving side removes cyclic prefix (CP) of areceived signal (75) and performs fast fourier transform (FFT) (76). Theresult is as expressed by the following Equation 14.

$\begin{matrix}{\begin{matrix}{{R(k)} = {{{C_{r}^{M}(k)}{H(k)}} + {N(k)}}} \\{= {{{{C_{guard}^{M}(k)} \cdot {\exp\left( {{- j}\frac{2\pi\; k\;\tau}{N_{fft}}} \right)}}{H(k)}} + {N(k)}}}\end{matrix}{k = 0},1,2,\ldots\mspace{14mu},{N_{fft} - 1}} & \left\lbrack {{Equation}\mspace{14mu} 14} \right\rbrack\end{matrix}$

The receiving side estimates the value of M of the GCL CAZAC codetransmitted from the transmitting side by using the FFT result (77). Thevalue of M can be estimated through the maximum value of IDFT value ofdifferential encoding in accordance with the following Equation 15.

$\begin{matrix}{M^{\prime} = {{Index}\left( {\max\limits_{n}\left( {{IDFT}\left( {{R\left( {k - 1} \right)}^{*}{R(k)}} \right)} \right)} \right)}} & \left\lbrack {{Equation}\mspace{14mu} 15} \right\rbrack\end{matrix}$

In the Equation 15 , when (R(k−1))*×R(k) is performed, a delay componentτ added intentionally by the transmitting side becomes a constant ofexp(−jτ) and thus does not affect estimation of the value of M. In theEquation 15 , ( )* means conjugate.

It will be apparent that maximum values are only detected duringdetection and some of the maximum values are selected as candidatevalues so as to process the selected values in parallel.

The receiving side detects the delay component τ after estimating thevalue of M (78). The delay component can be estimated through thefollowing steps.

In the first step, the received signal is compensated by the GCL CAZACsequence corresponding to M′ estimated by the Equation 15 , and thisstep can be performed by the following Equation 16.R′(k)=R(k)·(C ^(M)(k))*  [Equation 16]

Once the first step is performed, frequency response of the channel inthe frequency domain can be obtained.

In the second step, IFFT is performed for the resultant value of theEquation 16 , and this step can be performed by the following Equation17.r′(n)=IFFT(R′(k))  [Equation 17]

Once the second step is performed, impulse response in the time domaincan be obtained.

FIG. 10 is a view illustrating the resultant value after the second stepis performed.

In the third step, powers within each of delay groups are added, and thegroup having the greatest value of the added values is selected. Thisstep can be performed by the following Equation 18.

$\begin{matrix}{D^{\prime} = {{Indexofgroup}\left( {\max\limits_{g}\left( {\sum\limits_{l = 0}^{{{Nfft}\text{/}{Ng}} - 1}\;\left| {\left( r^{\prime} \right)^{g}\left( {{\frac{N_{ffl}}{N_{g}} \cdot g} + l} \right)} \right|^{2}} \right)} \right)}} & \left\lbrack {{Equation}\mspace{14mu} 18} \right\rbrack\end{matrix}$

In the Equation 18 , N_(g) means the number of groups that can generatedelay, and g means group indexes (0,1, . . . ).

FIG. 11 illustrates a performance curve according to the preferredembodiment of the present invention. Referring to FIG. 11, a detectionerror rate is shown under the experimental environment of the TU channelmodel of 3 km/h.

In FIG. 11, i) ‘Delayed S-SCH’ illustrates detection performance whichdetects the delay group only,

ii) ‘Only GCL(IDFT)’ illustrates a performance graph when the GCL codeaccording to the related art is only used, and

iii) ‘Proposed’ illustrates a performance graph from the methodaccording to the preferred embodiment of the present invention.

iii), which corresponds to the present invention, is obtained bycombination of i) and ii). Since suggested overall performance dependson poorer performance of i) and ii), it depends on performance of ii).Accordingly, it is noted that overall performance is obtained almostequally to that obtained by the related art only.

It is noted that although the number of available cell IDs in case ofN=75 is 40 in the related art, the number of cell IDs increases to40×8=320 without degradation of performance in the preferred embodimentof the present invention.

In the case that two base stations use the same M and different delayvalues in accordance with the present invention, a user equipmentdetects two delay groups when two signals simultaneously reach the userequipment. To avoid this specific circumstance, a limitation that twobase stations should use different values of M can be considered. Inthis case, although the number of cell IDs is equal to that of therelated art (40), information (8 types) by means of delay can be usedfor other secondary information (ex., frame synchronizationinformation).

In the hierarchical structure, initial synchronization is performed bythe P-SCH, and cell ID is detected through the S-SCH. On the other hand,in the non-hierarchical structure, both initial synchronization and cellID detection are performed through the SCH. Initial synchronization ofthe hierarchical structure is performed by cross correlation using acommon signal which each user equipment knows as every base stationtransmits the common signal through the P-SCH. On the other hand,initial synchronization of the non-hierarchical structure is performedby blind detection using auto correlation through a repeated structureof OFDM signals transmitted from the SCH. Accordingly, since there is noneed to know a specific signal, anything transmitted from the SCH canundergo synchronization. In this regard, in the non-hierarchicalstructure, cell ID information is transmitted along with the SCH. If thetechnical spirit according to the present invention is applied to thenon-hierarchical structure, cell ID information is transmitted andreceived in the same method as that applied to the S-SCH in thehierarchical structure.

According to the present invention, the following advantages can beobtained.

First, it is possible to extend the range of the code sequences that canbe used in the communication system.

Second, in case of the synchronization channels, secondary informationthat can be transmitted can be increased.

Third, it is possible to prevent codes from being decreased bymulti-path.

Finally, frequency diversity effect can be obtained when the receivingside performs code estimation.

It will be apparent to those skilled in the art that the presentinvention can be embodied in other specific forms without departing fromthe spirit and essential characteristics of the invention. Thus, theabove embodiments are to be considered in all respects as illustrativeand not restrictive. The scope of the invention should be determined byreasonable interpretation of the appended claims and all change whichcomes within the equivalent scope of the invention are included in thescope of the invention.

What is claimed is:
 1. A method for transmitting a signal by atransmitting side device, the method comprising: generating, by asequence generator in the transmitting side device, a sequence of afrequency domain with a Constant Amplitude Zero Auto-Correlation (CAZAC)property; mapping the sequence into subcarriers; performing an InverseFast Fourier transform (IFFT), by an IFFT processor in the transmittingside device, on the subcarriers to convert the mapped sequence into atime domain sequence; performing a cyclic shift, by a shifter in thetransmitting side device, on the time domain sequence; and transmitting,by a transmitter in the transmitting side device, the cyclically shiftedtime domain sequence to a receiving side device, wherein signalinginformation is signaled through the cyclically shifted time domainsequence generated by the cyclic shift, wherein the signalinginformation includes a length of a cyclic prefix, and wherein thesequence is generated based on a Zadoff-Chu sequence.
 2. A transmittingside device comprising: a processor to map a sequence of a frequencydomain with a Constant Amplitude Zero Auto-Correlation (CAZAC) propertyinto subcarriers and perform an Inverse Fast Fourier Transform (IFFT) onthe subcarriers to convert the mapped sequence into a time domainsequence, wherein a cyclic shift is performed on the time domainsequence; and a transmitter to transmit the cyclically shifted timedomain sequence to a receiving side device, wherein signalinginformation is signaled through the cyclically shifted time domainsequence generated by the cyclic shift, wherein the signalinginformation includes a length of a cyclic prefix, and wherein thesequence is generated based on a Zadoff-Chu sequence.